This invention relates to the field of switching DC-to-DC power conversion and in particular to the new class of switching converters employing novel type of lossless switching which provides simultaneously the ultra high efficiency in a very compact size and additional performance advantages, such as much reduced EMI noise and much improved reliability.
Definitions and Classifications
The following notation is consistently used throughout this text in order to facilitate easier delineation between various quantities:
1. DCxe2x80x94Shorthand notation historically referring to Direct Current but by now has acquired wider meaning and refers generically to circuits with DC quantities;
2. ACxe2x80x94Shorthand notation historically referring to Alternating Current but by now has acquired wider meaning and refers to all Alternating electrical quantities (current and voltage);
3. i1, v2xe2x80x94The instantaneous time domain quantities are marked with lower case letters, such as i1 and v2 for current and voltage;
4. I1, V2xe2x80x94The DC components of the instantaneous periodic time domain quantities are designated with corresponding capital letters, such as I1 and V2,
5. xcex94i1xe2x80x94The difference between instantaneous and DC components is designated with xcex94, hence xcex94i1 designates the ripple component or AC component of current i1;
6. Dxe2x80x94The duty ratio of the input switch S1 is defined as D=tON/TS where tON is ON time of the input switch, and TS is the switching period defined as TS=1/fS where fS is a constant switching frequency. Switch S1 is closed and conducts current (turned ON) during DTS interval;
7. Dxe2x80x2xe2x80x94The complementary duty ratio Dxe2x80x2 of the input switch S1 is defined as Dxe2x80x2=1xe2x88x92D and Dxe2x80x2TS is interval during which input switch S1 is open (turned OFF);
8. S1, S2, Sxe2x80x21, Sxe2x80x22xe2x80x94Switch designations respectively for input switch, output switch, complementary input switch, and complementary output switch and, at the same time, designate the switching states of the respective active, controllable switches as follows: high level indicates that active switch is turned ON, low (zero) level that it is turned OFF;
9. CR1xe2x80x94Designation for the current rectifier (CR) diode and its corresponding switching time diagram. Since diode is a two-terminal passive switch, switching time diagram represents also the state of diode switch as follows: high level indicates that the diode is ON and low level that it is OFF;
10. Ixe2x80x94One quadrant switch is designated by Roman number (I through IV) within a rectangular box around ideal switch signifying its limitation to particular one-quadrant operation;
11. CBSxe2x80x94Designates the Current Bi-directional Switch as a three-terminal controllable semiconductor switching device, which conducts the current in either direction in an ON-state, but blocks the voltage of only one polarity in an OFF-state between two power terminals and has a third controlling terminal to independently control the state of the switch between two power terminals;
12. VBSxe2x80x94Designates the Voltage Bi-directional Switch as a three-terminal controllable semiconductor switching device, which conducts the current in only one direction in the ON-state, but blocks the voltage of either polarity in an OFF-state between the two power terminals and has a third controlling terminal to independently control the state of the switch between two power terminals;
13. CBS/VBSxe2x80x94Designates that either Current Bi-directional Switch (CBS) or Voltage Bi-directional Switch (VBS) can be used.
The demand for reduced size and weight of electronic power processing equipment to make it compatible with ever shrinking size of electronic signal processing equipment resulted in the continuous push toward increasing the switching frequency at which DC-to-DC switching converters operate: from initial 20 kHz level to 200 kHz and higher switching frequencies. This, in turn, results in proportionally increased switching power losses. Hence, in the past, a number of converter topologies have emerged, which belong to two broad categories:
1. Hard-switching converter category in which no attempts were made to reduce the switching losses;
2. Soft-switching converter category in which measures were taken to reduce the switching losses.
Unfortunately, in most cases, the reduction of switching losses was accompanied with the increase of other losses, such as conduction losses of the switching devices or losses associated with energy stored in transformer leakage inductance and other additional losses, which resulted only in small to moderate improvements in efficiency.
The switching converters can also be classified into three classes relative to a number of switches employed:
1. Two-Switch Converter class, example of which is the prior-art buck converter.
2. Three-Switch Converter class such as prior-art forward converter;
3. Four-Switch Converter class such as the present invention and a number of other prior-art converters.
Prior-Art Soft-Switching Converters
One of the first soft-switching methods which provided reduction of switching losses was introduced by C. Henze, H. C. Martin and D. W. Parsley in xe2x80x9cZero-Voltage Switching in High-Frequency Power Converters Using Pulse-Width Modulationxe2x80x9d, IEEE Applied Power Electronics Conference, (IEEE Publication 88CH2504-9) pp33-40, 1988 record on a basic buck converter which belongs to Two-Switch Converter class and is shown in prior-art of FIGS. 1(a-g).
In order to obtain zero-voltage switching at a constant switching frequency, the usual transistor-diode implementation of two switches is replaced with two MOSFET transistors, each of which is modeled as a parallel connection of an ideal switch with an anti-parallel xe2x80x9cbodyxe2x80x9d diode and a parasitic drain-to-source capacitor, resulting in circuit models of FIGS. 1(c-f). The total switching cycle TS is broken down into 4 intervals by proper drive timing of the two switches S and Sxe2x80x2 as shown in FIG. 2e. Note that with two controllable switches, two well defined transition intervals are introduced during which both switches are OFF. The first transition interval (tN in FIG. 2e), starts when switch S is turned OFF (as in FIG. 2e) and is also known as the xe2x80x9cnaturalxe2x80x9d transition (DTS to Dxe2x80x2TS transition, or simply D to Dxe2x80x2 transition). By turning OFF the switch S, the inductor current IP is flowing naturally in a needed direction (represented by the current source IP on FIG. 2a-f). This current source IP charges the parasitic capacitor CS of switch S and discharges parasitic capacitor Cxe2x80x2S of switch Sxe2x80x2 until capacitor Cxe2x80x2S is fully discharged at which instant the body-diode of switch Sxe2x80x2 clamps the voltage at zero and prevents reverse charging of capacitor Cxe2x80x2S of switch Sxe2x80x2. At that instance, the switch Sxe2x80x2 can be turned ON with zero switching losses (FIG. 2e), since the charge of Cxe2x80x2S was already relocated to capacitance CS of the switch S (charged to Vg). In order to perform the reverse process during the Dxe2x80x2 to D transition, a negative inductor current IN is needed. The simplest method to accomplish this is to design the output inductor to have a large ripple current, such that its peak-to-peak ripple current is at least 3 times the maximum DC load current. As seen in the inductor current waveform in FIG. 2e, the instantaneous inductor current iL will at some point during Dxe2x80x2TS interval reverse direction and become negative with magnitude IN. Just before the end of complementary interval Dxe2x80x2TS the switch Sxe2x80x2 is turned OFF initiating the so-called xe2x80x9cforcedxe2x80x9d transition (since the inductor current is now intentionally forced to become negative by the converter circuit designed for large ripple). During this forced transition interval (tF in FIG. 2e), the opposite to tN interval occurs: this negative inductor current IN charges parasitic capacitor Cxe2x80x2S of switch Sxe2x80x2 and discharges parasitic capacitor CS of switch S until voltage VS of S reaches zero. At that instant body-diode clamps the voltage on switch S to zero forcing switch S to turn-ON at zero voltage in a lossless manner. Hence recycling of the charge stored in the parasitic capacitors CS and Cxe2x80x2S is provided instead of being dissipated each cycle as in xe2x80x9chard-switchingxe2x80x9d.
Even though soft-switching can be achieved on both active switches S and Sxe2x80x2 in this very simple manner, and the voltage stresses on the switches are low, the big disadvantage is that the magnitude of the output inductor ripple current must be more than two times higher than the maximum DC load current in order to achieve the soft-switching for all operating conditions. Clearly, this soft-switching method suffers from the need to have a large inductor ripple current so that a negative instantaneous inductor current is obtained before the end of Dxe2x80x2TS interval in order to accomplish the forced Dxe2x80x2 to D transition. This, in turn, increases the conduction losses significantly and thus diminishes to a large extent the savings obtained by reduced switching losses. In addition, an increased size of output capacitor is needed to absorb this large ripple current and to reduce the output AC ripple voltage to acceptable level.
Another prior-art method of reduction of switching losses belongs to the Three-Switch Converter class, as disclosed by U.S. Pat. No. 4,415,959 issued to P. Vinciarelli, for xe2x80x9cForward Converter Switching at Zero Currentxe2x80x9d. To force the main input power switch to switch at zero current in this quasi-resonant converter, the reactive components, small resonant inductor and small resonant capacitor are used to distort the main switch square-wave like current waveform into a sinusoidal-like current waveform. This makes possible turning ON and OFF of the main switch at zero current and reduces its switching losses caused by switch current and switch voltage overlap and by finite switching time characteristic of the semiconductor switching devices. Unfortunately, the increased RMS value of the switch current increases the conduction losses, thereby diminishing some of the switching loss reduction gained by zero current switching. More serious, however, is the fact that the dominant switching loss due to xc2xdCV2 energy stored on the parasitic capacitance of the main switch still remains and is dissipated when that switch is turned ON. This switching loss is especially pronounced in applications operating from high input DC voltages, such as nominal 300V DC input voltage in OFF-line applications, using rectified AC line as a DC source.
The converter disclosed in U.S. Pat. No. 4,441,146 issued to P. Vinnciarelli for xe2x80x9cOptimal resetting of the transformer""s core in single-ended forward convertersxe2x80x9d belongs to the of Four-Switch Converter class. The branch comprising the auxiliary switch and storage capacitance, and placed on transformer secondary was used with a sole purpose to form a xe2x80x9cmagnetizing current mirrorxe2x80x9d to reset the transformer""s magnetic core and has no other roles. On the contrary, in the present invention, the branch comprising an auxiliary switch and an auxiliary capacitor is placed on the primary side of the novel switching converter topology accomplishing not only the transformer""s magnetic core reset but also more importantly the elimination of switching losses.
The converter disclosed in the U.S. Pat. No. 5,291,382 issued to Isaac Cohen for xe2x80x9cPulse Width Modulated DC/DC Converter With Reduced Ripple Current Component Stress and Zero Voltage Switching Capabilityxe2x80x9d also belongs to the Four-Switch Converter class. In this converter, the soft-switching at zero voltage is achieved in a method analogous to the buck converter of FIGS. 1(a-g). It is based on the small magnetizing inductance of the isolation transformer which results in large magnetizing ripple current, hence with the same soft-switching and efficiency limitations as in a soft-switching buck converter. However, since soft-switching is accomplished by large magnetizing current ripple of transformer and not with a large output inductor ripple current as in a buck converter, the undesirable effect of large output inductor ripple current of the buck converter on output ripple voltage is eliminated.
Yet another example of the Four-Switch Converter class is the prior-art converter disclosed in the U.S. Pat. No. 5,066,900 issued to John Basset for xe2x80x9cDC/DC Converter Switching at Zero Voltagexe2x80x9d. In this converter, the leakage inductance of the transformer is used as a resonant inductor to force the reduction of switching losses, However, the use of the passive rectifier diodes for the two switches on the converter""s output (secondary side) instead of the controllable switches with optimum switching time control as in the present invention, severely limits the efficiency improvements which can be achieved with this soft-switching technique and especially so for the applications with moderate to high input DC voltages such as off-line converter applications.
The common to all above cited prior-art soft-switching converters is that although they employ different soft-switching methods on the members of Three-switch and Four-switch Converter class, they all utilize only the passive current rectifier switches for the two output switches. To the contrary, the present invention, which belongs to the Four-switch Converter class uses in addition to the two active switches on input side also two active and controllable switches on the output secondary side which are either of the CBS or VBS variety. Together with the very special switching time control of all four controllable switching devices this results in elimination of switching losses without any undesirable increase of other losses, such as conduction losses, leakage losses, etc., as was the case with the prior-art soft-switching methods. Thus, the present converter with its unique switching time control belongs to a new, third category of switching converters (in addition to hard-switching and soft-switching categories introduced earlier) characterized by near zero switching losses over the wide operating range, and therefore termed Lossless Switching Converter class.
From the above review, it is clear that a new lossless switching method is needed which eliminates the switching losses without introducing all the other undesirable features associated with classical prior-art soft-switching methods and thereby significantly increases the overall efficiency as confirmed both theoretically and on experimental prototypes. This invention introduces such novel lossless switching methods, which require in addition to suitable converter topology and proper semiconductor switch type also a proper drive timing of the four controllable switches and result in elimination of the switching losses. Furthermore, switching loss elimination is maintained in some cases throughout the whole duty ratio operating range.
A primary objective of this invention is to provide a lossless switching DC-to-DC converter, which, through the use of novel lossless switching methods, eliminates switching losses (without increasing other converter losses) to achieve ultra high efficiency heretofore unachievable by the prior-art soft-switching methods. The novel lossless switching allows operation at ultra high switching frequencies and thereby substantial reduction in size and weight and increased power density. The inherent additional benefits are reduced EMI noise and reduced component stress (voltage, current, and temperature) for increased reliability.
Lossless Switching Converter Categories
The new lossless switching DC-to-DC converter is comprised of a Power Processing Stage with up to four controllable switches with specific switching-quadrant characteristics depending on the switch location in the converter itself (for example current bi-directional or voltage bi-directional) and the Switching Time Control Box, which provides the needed sequence of switching via electronic drive control for at least three or for all four controllable switches to achieve lossless switching in a number of alternative ways. The invention is embodied in both non-isolated power stage and isolated power stage, with output switch implemented as either CBS/diode output switch or as VBS output switch.
The isolated Power Processing Stage is comprised of an isolation transformer, input and output inductors, series input capacitor, auxiliary capacitor and four controllable switches. The input inductor is connected in series with the DC source and provides the non-pulsating (continuous) input current, while the output inductor is connected in series with the DC load and provides non-pulsating (continuous) output current. The series input capacitor is connected in series with the input inductor and transformer primary. Input switch and complementary input switch are on the transformer""s primary side, while output switch and complementary output switch are on the transformer""s secondary side. The branch with auxiliary capacitor in series with the complementary input switch is positioned within converter in such a way to conduct a small AC ripple current only while the complementary input switch is closed. This AC ripple current together with the controllable output switch and the novel switching time control enables lossless switching operation, with efficiency and size performance not possible with prior-art soft-switching converters. This branch also insures that the isolation transformer is automatically volt-second balanced without a need for reset winding or other core reset means. There are many alternative ways to connect this branch to the remaining part of the new converter and still retain the above unique properties. A number of such possibilities are disclosed in detailed specifications along with the general functional criteria that such branch must satisfy. The non-isolated variant is obtained by simply replacing the isolation transformer with inductor.
Both isolated and non-isolated lossless switching converters can be implemented in two variants based on the type of the output controllable switch: CBS or VBS.
Lossless Switching With CBS Output Switch
The first lossless switching category with CBS output switch utilizes a resonance between the parasitic capacitances of the switches and the leakage inductance of the isolation transformer (or separate additional resonant inductor in case of non-isolated converter) to achieve lossless switching. The special switching time control of controllable switches gives rise to a very effective resonant current capable of producing lossless switching and providing the ultra high efficiency and small size heretofore unattainable by prior-art converters.
This category is broken down into three main subcategories as follows:
1. A non-isolated converter without resonant inductor with only switching time control adjustments of the CBS semiconductor switches, which provides a substantial switching loss reduction,
2. A non-isolated converter, which includes an additional resonant inductor and relies on its resonance with parasitic capacitances of the switches to accomplish lossless switching.
3. An isolated converter, whose leakage inductance plays the role of the resonant inductor and practically accomplishes elimination of all switching losses by use of the novel lossless switching time control of controllable CBS switches.
Lossless Switching With VBS Output Switch
The second lossless switching category with the VBS output switch does not depend on the resonant current as CBS output switch category did. Instead, it is based on the auxiliary capacitor AC ripple current and the voltage bidirectional property of the output switch to force the lossless discharge of the respective parasitic capacitances to zero voltage, at which point, appropriate controllable switches can be turned ON with zero losses. As this method does not depend on resonance, it is independent of the operating point and maintains such ideal switching performance over wide operating range including the whole input voltage range and for any load current from no load to full load.
This category is also broken down into three subcategories as follows:
1. A non-isolated switching converter without resonant inductor, which uses a very specific switching time control sequence to obtain lossless switching performance and results in symmetrical voltage waveform on the output switch, thus termed xe2x80x9csymmetricalxe2x80x9d lossless switching.
2. A non-isolated converter with resonant inductor, which uses another very specific switching time control sequence to obtain lossless switching performance and results in asymmetrical voltage waveform across the output switch, hence it is termed xe2x80x9casymmetricalxe2x80x9d lossless switching.
3. An isolated converter, which inherently includes the leakage inductance of the isolation transformer but does not depend on it for lossless switching performance, utilizes the novel switching time control analogous to the above xe2x80x9casymmetricalxe2x80x9d lossless switching time control.
Another converter improvement applicable to both CBS and VBS category is the one in which the input inductor and the isolation transformer are combined onto a common magnetic core to form an Integrated Magnetic circuit, which results in desirable zero-ripple input inductor current over the full operating range and thus reduces the conducted EMI noise as well as input ripple voltage. In non-isolated version, the input and middle inductors are combined into a Coupled-Inductor structure with the same benefits.
The novel features that are considered characteristic of this invention are set forth with particularity in the appended claims. The invention will best be understood from the following description when read in connection with the accompanying drawings.